FIG. 1(a) depicts a multipath channel through which television signals propagate from a transmitter 1 to a receiver 2. As depicted, the television signals arrive at the receiver 2 via a number of paths A, B, C, D including a short direct path A, and longer paths B, C, D in which the signals reflect off of features of the channel (e.g., buildings, mountains, and the ionosphere). All of these signals are superimposed at the receiver 2. The signals arriving via the paths B-D are weaker than the signal arriving via the direct path A. Thus, the signal arriving via the path A produces the strongest video image at the receiver 2 and is referred to as the "main" signal. Furthermore, the signals arriving via the paths B-D are delayed with respect to the main signal arriving via the path A. As a result, the signals arriving via the paths B-D produce delayed duplicate video images or "post-ghosts" of the main signal arriving via the path A as depicted in FIG 1(b).
Another multipath channel is depicted in FIG 1(c). As depicted, a signal arrives via a short path E through buildings 3. A signal also arrives via a longer reflection path F and is thus delayed with respect to the signal arriving via the short path E. In this case, it is assumed that the signal arriving via path E is attenuated to a greater extent (by virtue of propagating through the buildings 3) than the signal arriving via the path F. In such a case, the weaker signal arriving via the path E produces a "pre-ghost" of the main signal arriving via the path F as depicted in FIG 1(d).
It is desirable to eliminate both pre-ghosts and post-ghosts of the main signal in order to improve reception. Several ghost cancelling systems have been proposed in the form of a channel equalizer. FIG. 2(a) depicts a transmission path including a transmitter 4, a multipath channel 5 and a receiver 6 which includes a channel equalizer 7 and a display device 8. In such systems, an ideal ghost cancelling reference (GCR) signal R.sub.ideal (t) is inserted into the video signal V(t), e.g., during the vertical blanking interval, prior to transmission from the transmitter 4. The transmitter 4 transmits the video signal V(t) (including the ideal GCR signal R.sub.ideal (t)) which propagates through the multipath channel 5 having an impulse response A(t). By virtue of propagating through the multipath channel 5, a signal with ghosts V(t)*A(t) (including R.sub.ideal (t)*A(t)) is produced, where "*" means "convolved with". This signal V(t)*A(t) is received at the receiver 6 where it is inputted to the channel equalizer 7. The channel equalizer 7 has an impulse response W(t) and therefore outputs the signal V(t)*A(t)*W(t). The channel equalizer 7 is designed so that V(t)*A(t)*W(t)=V(t). The signal outputted from the channel equalizer 7 is then displayed on a display device such as a cathode ray tube (CRT) screen 8.
The channel equalizer 7 is shown in greater detail in FIG. 2(b). As depicted in FIG. 2(b), the channel equalizer 7 typically includes an analog to digital converter (ADC) 9 which converts the received video signal V(t)*A(t) to digital form. Illustratively, the received video signal V(t)*A(t) has an upper cutoff frequency of approximately 4.2 MHz. The received video signal V(t)*A(t) is illustratively sampled in the ADC 9 at 14.32 MHz. These samples are inputted to an extraction circuit 10 which extracts the received GCR signal R.sub.rec (t) (where R.sub.rec (t)=R.sub.ideal (t)*A(t)) from the received video signal V(t)*A(t). This received GCR signal R.sub.rec (t) may be temporarily stored in a RAM 11. The received GCR signal R.sub.rec (t) is then compared to an ideal GCR signal R.sub.ideal (t) (obtained from a circuit 12, such as a ROM) in a CPU or digital signal processor (DSP) 13. Based on the discrepancy between the received R.sub.rec (t) and the ideal R.sub.ideal (t) GCR signals, the CPU or DSP 13 generates filtering or tap coefficients for cancelling ghosts in the received video signal V(t)*A(t). The tap coefficients are transferred to a transversal filter 14. The received video signal V(t)*A(t) is accordingly digitally filtered by the transversal filter 14 using the tap coefficients determined by the CPU or DSP 13. The filtered video signal outputted by the transversal filter 14 may illustratively be converted back to analog form in a digital to analog converter (DAC) 15.
FIG. 2(c) shows an exemplary prior art transversal filter 14 including a finite impulse response filter (FIR) 16, and an infinite impulse response filter (IIR) 17. Illustratively, the IIR filter 17 is formed by connecting a second FIR filter 18 in negative a feedback path of an adder 19 to which the FIR filter 16 is connected.
Several conventional algorithms have been proposed for obtaining FIR and IIR tap coefficients (see U.S. Pat. No. 4,947,252). Tap coefficients of the transversal filter 14 (FIG. 2(c)) within the channel equalizer 7 (FIG. 2(a)) may be derived from the formula V(t)*A(t)*W(t)=V(t). According to one conventional method, called a division method, the tap coefficients are determined so that ##EQU1## where V(f), A(f) and W(f) are the video signal, the channel impulse response and the equalizer impulse response, in the frequency domain, respectively. The video signal V(f) and the channel impulse response A(f) are both unknown and vary over time. If, however, a known ghost cancelling reference (GCR) signal R.sub.ideal (t) is inserted into the video signal V(t) prior to transmission, then the tap coefficients may be generated by comparing the received and ideal GCR signals R.sub.rec (t) and R.sub.ideal (t). In such a case, the above formula may be simplified to: ##EQU2##
FIG. 3 depicts one conventional division method algorithm for obtaining tap coefficients. Typically, the FIR filter 16 utilizes a relatively small number of taps for cancelling "nearby" ghosts (e.g., a ghost separated by 2 .mu.sec from the main signal). To determine nearby ghost tap coefficients according to this method, the signal R.sub.rec (t) is first windowed over a short interval appropriate for cancelling nearby ghosts to produce the signal R.sub.rec '(t) in step 340 (herein, one prime mark indicates short term windowing). Next, in step 342, the signal R.sub.rec '(t) is fourier transformed to produce the signal R.sub.rec '(f). Then, in step 344, the nearby ghost tap coefficient signal W.sub.near (f) is determined by the formula ##EQU3##
In step 346, this signal W.sub.near (f) is converted to the time domain by computing its inverse fourier transform. Finally, in step 348, the signal W.sub.near (t) is windowed over a short interval (appropriate for producing nearby ghost tap coefficients) to produce the signal W.sub.near '(t).
The prior art division method of FIG. 3 also produces tap coefficients for the IIR filter 17. The IIR filter 17 typically has a large number of tap coefficients for cancelling "non-nearby" or "normal" ghosts (e.g., a ghost separated by 40 .mu.sec from the main signal).
In step 350, the received GCR signal R.sub.rec (t) is windowed appropriately for cancelling normal ghosts to produce the signal R.sub.rec "(t) (herein, two prime marks means long term windowing). Next, in step 352, the signal R.sub.rec "(t) is fourier transformed to produce the signal R.sub.rec "(f). In step 354, the windowed nearby tap coefficient signal W.sub.near '(t) (obtained in step 348) is fourier transformed to produce the signal W.sub.near '(f). These two signals R.sub.rec "(f) and W.sub.near '(f) are used to form the signal h(f) in step 356. h(f) is determined by the formula: EQU h(f)=R.sub.ideal (f)-R.sub.rec "(f).cndot.W.sub.near '(f) (4)
Then in step 358, the normal ghost tap coefficient signal W.sub.norm (f) is determined by the formula: ##EQU4## In step 360, the inverse fourier transform of W.sub.norm (f) is computed to produce the signal W.sub.norm (t). Finally, W.sub.norm (t) is windowed over a long interval (appropriate for producing normal ghost tap coefficients) to produce the signal W.sub.norm "(t) in step 362.
Several different circuits have been proposed for implementing the FIR or IIR filters 16 or 17 (see U.S. Pat. No. 4,953,026). FIG. 4 shows a first prior art circuit 40, suitable for implementing a FIR filter 16, called a one-tap selection circuit. Digital samples of a received video signal which are inputted to the FIR filter 40 are shifted into a shift register 42. These digital samples are read out of the shift register 42 by a tap selection circuit 43. In addition, a plurality of tap coefficients (determined, for example, according to the aforementioned division method) are inputted to a tap gain controller circuit 46. The tap gain controller circuit 46 outputs the largest tap coefficient from the tap coefficients inputted thereto to a plurality of multipliers 44. The tap gain control circuit also causes the tap selection circuit 43 to supply selected video samples from the shift register 42 to particular multipliers 44. Each supplied video sample is multiplied in the multipliers 44 by the largest tap coefficient (which tap coefficient is supplied by the tap gain controller circuit 46). These products are then added together in an adder 45.
FIG. 5 shows a second prior art circuit 50, suitable for implementing a FIR filter, called a one-tap variable delay circuit. In this circuit 50, the video samples are inputted, in parallel, to a plurality of variable delay elements 52. A tap gain control circuit 55 controls the delay period of each variable delay element 52 and also outputs a single tap coefficient to each of the multipliers 53. The video samples are outputted from each variable delay element 52 to the multipliers 53 where they are multiplied by the tap coefficient supplied by the tap gain control circuit 55. The products outputted by the multipliers 53 are then added together in an adder circuit 54.
A third prior art circuit 60, called a multi-tap selection circuit, is shown in FIG. 6. Video samples are shifted into a shift register 62. A highly complex tap selection circuit 63 is provided for selecting groups of video samples in the shift register 62 for output to transversal filters elements 68. A prior art transversal filter element 68 is shown in greater detail in FIG. 6(a). Video samples are received in a shift register 64 comprising a plurality of one sample delays 65. The video samples are supplied from each one of the sample delays 65 of the shift register 64 to corresponding multipliers 67. Each of the multipliers 67 also receives a tap coefficient of an array of tap coefficients and multiplies the video samples inputted thereto by this tap coefficient of the array. The products outputted by the multipliers 67 are added together in an adder 61. The outputs of the transversal filter elements 68 (FIG. 6), in turn, are added together in an adder circuit 66 (FIG. 6). As depicted in FIG. 6, the circuit 60 also has an output 69 for shifting out video samples which propagate through the shift register 62. This output 69 may be connected to the shift register of another similar circuit 60 connected sequentially to the first circuit 60.
Finally, U.S. Pat. No. 4,953,026 discloses a fourth prior art circuit 20 or 28 (FIGS. 7(a)-(b)), called a multi-tap variable delay circuit. In FIG. 7(a), video samples are received, in parallel, into a plurality of variable delay elements 22. The delay period of the variable delay elements 22 is controlled by a tap gain control circuit 25. Each variable delay element 22 outputs the video samples received therein to a corresponding transversal filter element 23 where the video samples are multiplied by an array of tap coefficients. As with the prior art circuits 40 (FIG. 4) and 50 (FIG. 5), the tap coefficients are supplied by the tap gain control circuit 25. The outputs of the transversal filter elements are then added together in an adder 24.
FIG. 7(b) shows an alternative embodiment of the multi-tap variable delay circuit 28 in which the variable delay elements 26 are daisy chained, i.e., sequentially connected in tandem. Video samples are inputted to the first variable delay element 26-1 and propagate therethrough. The video samples are then outputted to the second variable delay element 26-2 and from the second variable delay element 26-2 to the third variable delay element 26-3, etc. The circuit 28 is otherwise similar to the circuit 20 (FIG. 7(a)).
Each of these prior art circuits has disadvantages. The one tap selection circuit 40 (FIG. 4) requires a tap selection circuit 43 (FIG. 4) which is extremely complex and requires a large number of components. Both the complexity and number of components increase when such circuits 40 (FIG. 4) are used to form non-nearby ghost cancelling IIR filters which typically require many tap coefficients. Thus, such a circuit 40 (FIG. 4) is both difficult to produce within an integrated circuit chip and very expensive to manufacture. The one tap variable delay circuit 50 (FIG. 5) does not use complex selector circuits. However, the number of variable delay elements 52 (FIG. 5) necessary for ghost cancelling, particularly, non-nearby ghost cancelling in IIR filters, is large. Thus, this circuit 50 (FIG. 5) is also both expensive to manufacture and difficult to produce within an integrated circuit chip.
The multi-tap selection circuit 60 (FIG. 6) and multi-tap variable delay circuit 20 or 28 (FIGS. 7(a)-(b)) are better suited for IIR filters. However, as with the one tap selection circuit 40 (FIG. 4), the multi-tap selection circuit 60 (FIG. 6) requires an overly complex tap selection circuit 63 (FIG. 6) which is expensive and difficult to implement on an integrated circuit chip. The multi-tap variable delay circuit 20 or 28 (FIGS. 7(a)-(b)) does not require an overly large selection circuit or variable delay circuit. This circuit 20 or 28 (FIGS. 7 (a)-(b)), however, is disadvantageous in practice because transversal filter elements 23 (FIGS. 7(b)) are wasted. This may be better appreciated by an example. Suppose the maximum delay period of each variable delay element 22 of FIG. 7(a) and of FIG. 7(b) is two hundred clock cycles. Furthermore, suppose it is desirable to cancel a first ghost which is delayed from the main video signal by fifty clock cycles and a second ghost which is delayed from the main video signal by three hundred fifty clock cycles. FIG. 8 is a graph showing the relationship of the ghosts and the main video signal.
Using the circuit 28 of FIG. 7(b), the first variable delay element 26-1 may be adjusted to a delay of approximately fifty clock cycles, the second variable delay element 26-2 may be adjusted to a delay of two hundred clock cycles and the third variable delay element 26-2 may be adjusted to a delay of approximately one hundred clock cycles. (The circuit 20 of FIG. 7(a) cannot cancel the second ghost because it cannot delay the video signal by three hundred fifty clock cycles). The first transversal filter element 23-1 is then used to cancel the first ghost and the third transversal filter element 23-3 is then used to cancel the second ghost. It may be appreciated, however, that the second transversal filter element 23-2 cannot be used at all and is thus wasted.
It is therefore an object of the present invention to provide a ghost cancelling circuit which overcomes the disadvantages of the prior art circuits.